System and method for dynamically biasing oscillators for optimum phase noise

ABSTRACT

Systems and methods for biasing frequency oscillators to minimize phase noise are disclosed. The system may comprise a tank circuit having an inductor, at least a first coupling capacitor and a second coupling capacitor. The system may further comprise a varactor circuit electrically connected to the first coupling capacitor and the second coupling capacitor. The system may further comprise at least one first metal oxide semiconductor (MOS) device electrically connected in shunt with the tank circuit and a bias voltage. The at least one first MOS device may be electrically connected to a first gate bias voltage configured to bias the at least one first MOS device such that a first gate-to-source voltage of the at least one first MOS device remains below the first threshold voltage.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Application No.62/101,795, filed Jan. 9, 2015, entitled “SYSTEM AND METHOD FORDYNAMICALLY BIASING OSCILLATORS FOR OPTIMUM PHASE NOISE,” which ishereby incorporated by reference in its entirety.

BACKGROUND

1. Technological Field

This disclosure is related to voltage controlled oscillators (VCO) anddigitally controlled oscillators (DCO). More particularly, thisdisclosure presents a system and method for biasing oscillators tominimize phase noise.

2. Background

Digitally controlled or voltage controlled oscillators may have verysensitive nodes where thermal noise may be converted to phase noisearound the oscillation frequency. Biasing these nodes may negativelyaffect the oscillator by lowering the Q-factor of the tank circuit thusincreasing its phase noise. Some systems can implement resistivecircuitry to minimize varactor noise contribution.

SUMMARY

In general, this disclosure describes techniques related to minimizingphase noise and jitter in frequency oscillators. The systems, methodsand devices of this disclosure each have several innovative aspects, nosingle one of which is solely responsible for the desirable attributesdisclosed herein.

One aspect of the disclosure provides a frequency oscillator. Thefrequency oscillator can have a tank circuit having an inductor. Thetank circuit can have a first coupling capacitor, and a second couplingcapacitor. The frequency oscillator can also have a varactor circuitelectrically coupled to the first coupling capacitor and the secondcoupling capacitor. The frequency oscillator can also have a first metaloxide semiconductor (MOS) device having a first gate, a first drain, anda first source. The first source can be electrically coupled to thevaractor circuit. The frequency oscillator can also have a second MOSdevice having a second gate, a second drain, and a second source. Thesecond source can be electrically coupled to the varactor circuitopposite the first source. The frequency oscillator can also have afirst input electrically coupled to the first drain and the second drainto receive a first bias voltage. The frequency oscillator can also havea second input electrically coupled to the first gate and the secondgate to receive a first gate bias voltage.

Another aspect of the disclosure provides a frequency oscillator havinga tank circuit. The tank circuit can have at least one inductor and atleast one capacitor. The tank circuit can also be electrically coupledin parallel to a variable capacitance circuit. The frequency oscillatorcan also have a first MOS device having a first gate, a first source,and a first drain. The first source can be electrically coupled to thetank circuit and the variable capacitance circuit. The frequencyoscillator can also have a second MOS device having a second gate, asecond source, and a second drain. The second source can be electricallycoupled to the tank circuit and the variable capacitance circuit. Thefrequency oscillator can also have a first input electrically coupled tothe first drain and the second drain, and configured to receive a firstbias voltage. The frequency oscillator can also have a second inputelectrically coupled to the first gate and the second gate. The secondinput can be configured to receive a first gate bias voltage. The firstgate bias voltage can be selected to bias the first MOS device such thata first gate-to-source voltage of the first MOS device remains below afirst threshold voltage, and to bias the second MOS device such that asecond gate-to-source voltage of the second MOS device remains below asecond threshold voltage, when the frequency oscillator is in operation.

Another aspect of the disclosure provides a method for biasing anoscillator circuit. The method can comprise generating an oscillatingoutput using a tank circuit electrically coupled to a varactor circuit.The method can also comprise biasing the varactor circuit using a firstMOS device and a second MOS device. The varactor circuit can beelectrically coupled to a first source of the first MOS device and to asecond source of the second MOS device. The method can also comprisebiasing the first MOS device and the second MOS device with a first gatebias voltage at a first gate of the first MOS device and at a secondgate of the second MOS device. The method can also comprise controllinga first transconductance of the first MOS device and a secondtransconductance of the second MOS device with the first bias voltageand the first gate bias voltage.

Another aspect of the disclosure provides an apparatus for producing anoscillating frequency. The apparatus can have a resonating means forstoring energy at a resonant frequency. The resonating means can haveand at least one inductor and at least one capacitor. The apparatus canalso have a variable capacitance means having a first end and a secondend, the first end and the second end being electrically coupled to theresonating means. The apparatus can also have a first transistor meanshaving a first gate, a first drain, and a first source. The first sourcecan be electrically coupled to the first end. The apparatus can alsohave a second transistor means having a second gate, a second drain, anda second source. The second source can be electrically coupled to thesecond end. The apparatus can also have a first biasing meanselectrically coupled to the first drain and the second drain. Theapparatus can also have a second biasing means electrically coupled tothe first gate and the second gate.

Other features and advantages of the present disclosure should beapparent from the following description which illustrates, by way ofexample, aspects of the disclosure.

DESCRIPTION OF THE DRAWINGS

The details of embodiments of the present disclosure, both as to theirstructure and operation, may be gleaned in part by study of theaccompanying drawings, in which like reference numerals refer to likeparts, and in which:

FIG. 1 is a schematic diagram of an oscillator circuit;

FIG. 2 is a schematic diagram of an oscillator circuit incorporatingtransistors to reduce phase noise;

FIG. 3 is a plot of gate-to-source voltage over time of the MOS devicesof FIG. 2;

FIG. 4A is a circuit diagram of another embodiment of an oscillatorcircuit;

FIG. 4B is comparison of four plot diagrams of voltage, current, andimpedance values of the oscillator circuit of FIG. 4A over time;

FIG. 5A is a circuit diagram of another embodiment of an oscillatorcircuit for FIG. 2; and

FIG. 5B is a comparison of four plot diagrams of voltage, current, andimpedance values of the oscillator circuit of FIG. 5A over time.

DETAILED DESCRIPTION

The detailed description set forth below, in connection with theaccompanying drawings, is intended as a description of variousembodiments and is not intended to represent the only embodiments inwhich the disclosure may be practiced. The detailed description includesspecific details for the purpose of providing a thorough understandingof the embodiments. In some instances, well-known structures andcomponents are shown in simplified form for brevity of description. Asused herein, like reference numerals refer to like features throughoutthe written description.

FIG. 1 is a circuit diagram of an oscillator circuit. An oscillatorcircuit (oscillator) 100 is shown. The oscillator 100 may have aninductor L₁ 102 electrically coupled to a pair of coupling capacitors C₁104 and C₂ 106. In some embodiments, the coupling capacitors C₁ 104 andC₂ 106 electrically “couple” the inductor L₁ 102 to a varactor circuit,shown as a pair of varactors 112. The varactors 112 may operate asvoltage controlled capacitors. The inductor L₁ 102 may further beelectrically coupled to a pair of coarse tuning capacitors: a coarsetuning capacitor C₃ 114 and a coarse tuning capacitor C₄ 116 (referredto hereinafter as “coarse capacitors”). The combination of L₁ 102, C₁104, C₂ 106, and the varactors 112 may also be referred to herein as a“tank circuit,” “tank” or “LC circuit” 110 (indicated in dashed lines).In some embodiments, the tank 110 can act as an electrical resonator,storing energy oscillating at a characteristic resonant frequency of thecircuit.

In some embodiments, the tank 110 may not have all of the componentsidentified in FIG. 1 and described herein, depending on theconfiguration of the oscillator 100. The tank 110 may have only a singlecapacitor and a single inductor. In some embodiments the tank 110 caninclude the L₁ 102 in combination with the C₁ 104, and the C₂ 106,further including the C₃ 114 and the C₄ 116. In another embodiment, thetank 110 can be considered to include the L₁ 102 in combination with theC₁ 104, the C₂ 106, the C₃ 114, and the C₄ 116, and the varactors 112.In yet another embodiment, the tank 110 may have additional inductive,capacitive, and resistive circuitry implemented to change or regulatethe resonant frequency of the tank 110.

The oscillator 100 can further have an input V_(bias) 120 (hereinafter“V_(bias) 120”). As used herein, an input may generally refer to anelectrical coupling that can receive a voltage input, for example. TheV_(bias) 120 may be or receive a variable or constant direct current(DC) voltage applied to C₁ 104 and C₂ 106 of the tank 110. Theoscillator 100 may further have an input V_(tune) 130. The V_(tune) 130may also be or receive a variable or constant DC voltage applied to thevaractors 112 to tune or adjust the frequency of the oscillator 100. Thecapacitance of the varactors 112 can further be a function of thedifference between the V_(bias) 120 and the V_(tune) 130. Additionally,as the capacitance of the varactors 112 is changed, the frequency of theoscillator 100 also changes.

In an embodiment, the input V_(bias) 120 and the input V_(tune) 130 incombination with the tank 110 and the varactors 112 can be combined asthe oscillator 100 configured as a VCO.

In some embodiments, the oscillator 100 may further comprise a bandcontrol input 140 (hereinafter, “band control 140”) electrically coupledto the coarse capacitors 114, 116. The coarse capacitors C₃ 114 and C₄116 may form a portion of a switchable capacitor array for use in theoscillator 100 (e.g., a VCO or a DCO). The coarse capacitors C₃ 114 andC₄ 116, in conjunction with a metal oxide semiconductor (MOS) device 146(hereinafter referred to as “switch 146”) and an inverter 144 cancomprise such a switchable array. The switch 146 can be n-type MOS(NMOS) or p-type MOS (PMOS) transistor, having a drain coupled thecoarse capacitor C₃ 114, a source coupled to the other coarse capacitorsC₄ 116, and a gate coupled to a control signal, shown as the bandcontrol 140. The band control 140 can be coupled to the switch 146 viaan inverter 144. In some embodiments, the inverter 144 may not bepresent.

The band control 140 can provide a digital control signal for a DCO togenerate a clock signal, for example. Thus the bottom portion of theoscillator 100 of FIG. 1, taken by itself, may comprise a DCO. The bandcontrol 140 may supply the digital signal to the inverter 144 and theswitch 146 to induce a voltage to the tank 110. The inverter 144 and theswitch 146 may not be required for certain VCO implementations.

In some embodiments, a number of resistors may be included in theoscillator 100 to minimize the phase noise contributed by the varactors112. For example, resistors 122 a, 122 b (collectively resistors 122)may be incorporated in the oscillator 100 in shunt between the V_(bias)120 and the tank 110. The value of the resistors 122 can be selected tobe high enough to minimize the noise contribution of the varactors 112but low enough so as to not increase the thermal noise of contributionof the tank circuit (that would negatively affect the Q-factor of thetank 110). Accordingly, the resistor value may be in the kilo ohm (kΩ)range (e.g., 1 kΩ-1000 kΩ). However, in such an embodiment, theresistors 122 themselves can contribute varying levels of thermal noise,resulting in phase noise and jitter induced in the output of theoscillator 100.

In a similar fashion, resistors 142 a, 142 b (collectively resistors142) may also be incorporated in the oscillator 100 to bias the switch146. Using the threshold voltage (V_(th)) of the switch 146, the valueof the resistors 142 can be selected to turn the switch 146 on and offbased on the signal supplied by the band control 140. The switch 146 canthus be used to reliably switch the coarse capacitors C₃ 114 and C₄ 116on or off from (or in and out of) the LC tank 110 so that the oscillator100 can precisely generate a specified high frequency signal accordingto the band control 140 signal. In some embodiments, the band control140 is a digital signal that will alternatively switch the coarsecapacitors C₃ 114 and C₄ 116 in and out of the oscillator 100 circuit tocreate the output signal (e.g., a clock signal).

In some embodiments, the thermal noise added by the resistors 122, 142may be a result of thermal flux inside the individual resistors 122,142. The thermal flux may add a noise component to the VCO frequency andultimately manifest as phase noise (in a VCO, for example) or jitter (ina DCO, for example). The phase noise/jitter degrades the spectral purityof the oscillator 100 and may negatively affect the Q-factor.

FIG. 2 is a circuit diagram of an oscillator circuit incorporatingtransistors to reduce phase noise. As shown, an oscillator 200 mayresemble the oscillator 100 and comprise several similar components.Accordingly, like numbers indicate like components.

The oscillator 200 may have the tank 110 as described above with theassociated varactors 112. The oscillator 200 may be configured toproduce an output voltage V_(tank) 220 at a specific or variableresonant frequency, similar to the oscillator 100. The V_(tank) is shownas V_(tank) _(_) ₁ 220 a and V_(tank) _(_) ₂ 220 b representing outputsof the oscillator 200. The outputs at V_(tank) _(_) ₁ 220 a and V_(tank)_(_) ₂ 220 b can be located at opposite ends (e.g., a first end and asecond end) of the varactors 112 or other variable capacitance circuit.

The oscillator 200 may further comprise transistors in place of theresistors 122, 142 of FIG. 1. As noted above, high value or highresistance/impedance resistors (e.g., the resistors 122, 142 of theoscillator 100), add phase noise or jitter to the output of theoscillator 100. This can manifest as a reduction in the Q-factor of theoscillator 100 or a disruption of its spectral purity. The replacementof such high value resistors with for example, transistors, can presenthigh impedance without contributing to the phase noise or jitterassociated with the resistors 122.

In some embodiments, the oscillator 200 includes MOS devices 202 a, 202b (collectively MOS devices 202) in place of the resistors 122 ofFIG. 1. The MOS devices 202 can be implemented as NMOS or PMOS devices,depending on requirements and polarity within the desired oscillatorcircuit. For example, the MOS devices can be MOSFETs (MOS Field EffectTransistors) or other transistors having similar characteristics. Thegates of the MOS devices 202 can be electrically coupled to an input toreceive a gate bias voltage V_(g) _(_) _(bias) 210. The V_(g) _(_)_(bias) 210 may be controlled to bias the operation of the MOS devices202 in order to control the impedance and transconductance of the MOSdevices 202. Such biasing can prevent the gate-to-source voltage(V_(gs)) of the MOS devices 202 from exceeding their respectivethreshold voltages. The characteristic threshold voltages of therespective MOS device 202, 204, may be referred to herein as “V_(th).”This is discussed in more detail below.

In some embodiments, the oscillator 200 may further have MOS devices 204a, 204 b (collectively, MOS devices 204). The MOS devices 204 may beelectrically coupled to the oscillator 200 in place of the resistors 142(FIG. 1). The gates of each of the MOS devices 204 may further beelectrically coupled to an input to receive a second gate bias voltage,V_(g) _(_) _(bias2) 222. The V_(g) _(_) _(bias2) 222 can be set to biasthe operation of the MOS devices 204. Accordingly, MOS devices 204 canprovide high impedance (from gate to source) and low noise by preventingthe V_(gs) of the MOS devices 204 from exceeding V_(th), similar toabove.

In some embodiments, the input for V_(g) _(_) _(bias) 210 may beelectrically connected to the gates of each of the MOS devices 202 tobias the transconductance of the MOS devices 202. As used herein,“transconductance” may generally refer to the current induced within theMOS devices 202 from their respective sink to drain. Such MOS devices202 may be implemented to bias the high swing nodes of the oscillator200.

The V_(g) _(_) _(bias) 210 may be set such that V_(g) _(_) _(bias) 210is less than or equal to the sum of the V_(bias) and the V_(th) of theMOS device 202. In other words:V _(g) _(_) _(bias) ≦V _(bias) +V _(th)  (1)

As used herein, the V_(th) can generally describe the minimumgate-to-source voltage (V_(gs)) differential required to induce aconductive path (transconductance) between the source and drain of theparticular MOS device 202. Thus, where V_(gs) exceeds V_(th), aconductive channel is gradually opened between the respective source anddrain, increasing transconductance of the given MOS device (e.g., theMOS devices 202, 204). In some embodiments, biasing the MOS devices 202,204 in such a manner may result in noise near oscillation peaks, due tothe increased transconductance and the peak V_(gs). However, theoscillator 200 is less sensitive to noise (e.g., phase noise, or jitter)at the oscillation peaks and any noise induced at the peaks of theoscillation may be negligible.

As the gate-to-source voltage (V_(gs)) of the MOS devices 202 (or theMOS devices 204) approaches or exceeds V_(th), the respectivetransconductance increases. Additionally, as the source-to-drainconductive channel is opened in the MOS devices 202, thermal noise mayalso increase. Increased V_(gs) may result in increased noise. Thisproduces a similar effect as the resistors 122. Increasedtransconductance can also decrease the impedance of the MOS device 202from source to drain, further increasing noise. Conversely, whentransconductance is very low, the impedance of the MOS device 202 isvery high and the conductive channel from source to drain is closed. Insome embodiments, the impedance may increase into the range of mega ohms(MΩ) under such circumstances. Furthermore, when the transconductance isvery low, the thermal noise added by the MOS devices 202 is also verylow.

Accordingly, when the MOS devices 202 are biased according to equation 1above, V_(gs) does not exceed the V_(th) of the MOS devices 202. Biasingthe varactors 112 in such a manner presents the high impedance that canprevent reduction of the Q factor, while minimizing the thermal noisethat would otherwise be present with use of the resistors 122 (FIG. 1).Therefore, use of the MOS devices 202 results in low noise yet highimpedance of the MOS devices 202 at the oscillation nodes.

In other embodiments, the V_(g) _(_) _(bias2) 222 may be adjusted in asimilar fashion. In use with a DCO, the resistors 142 (FIG. 1) may bereplaced with the MOS devices 204. When the gates of the MOS devices 204are appropriately biased to maintain a V_(gs) less than V_(th), inaccordance with equation 1 above, the impedance of the MOS devices 204remains high, reducing the amount of noise induced in the system in thesame manner as the MOS devices 202.

As noted above, tunable LC-based DCOs and many VCOs can have aswitchable capacitor array (e.g., the coarse capacitors C₃ 114 and C₄116) used for tuning the frequency produced by the oscillator 100, 200.When the coarse capacitors C₃ 114 and C₄ 116 are switched into thecircuit, their Q is limited by the resistance of the switch 146, whichif too low, can reduce the Q of the tank leading to higher inducedthermal noise. When the capacitors C₃ 114 and C₄ 116 are switched off,or out of the circuit, the MOS devices 204 can be used to bias thesource and drain of the switch 146 to ensure the switch 146 turns offwith a reasonable settling time. In use with the oscillator 100, theresistors 142 need to be large so as to not degrade the Q of the tank110, so they can take up a relatively large physical area. Conversely,the use of the MOS devices 204 presents high impedance without theassociated noise of the resistors 142. In some embodiments, the MOSdevices 204 also occupy less space than a resistor 142 with comparableimpedance.

FIG. 3 is a plot of gate-to-source voltage over time of the MOS devicesof FIG. 2. A plot 300 depicts V_(gs) on the vertical (y) axis versustime (t) on the horizontal (x) axis. The V_(gs) on the y axis isrepresentative of the V_(gs) of an exemplary MOS device implemented asthe MOS devices 202. The V_(gs) of the plot 300 is also representativeof the MOS devices 204 when the coarse capacitors C3 114, C4 116 areswitched on and off. In operation, one of the coarse capacitors C3 114,C4 116 will be switched off when the other is switched on, resulting ina voltage response similar to the plot 300. In such an embodiment, theoutput may be a square wave or other variety of clock signal. The plot300 further depicts the V_(th) of the MOS devices 202, 204 as a dashedline 302.

In some embodiments, the plot 300 is a graphical representation of theoscillation frequency (e.g., a sine wave) of an associated VCO (e.g.,the oscillator 200) as biased by the gate voltage V_(g) _(_) _(bias) 210of FIG. 2. The plot 300 may also be representative of the DCO (e.g., theoscillator 200) biased by the gate voltage V_(g) _(_) _(bias2) 222. Inan embodiment, the amplitude (in volts) of the oscillations ismaintained below the V_(th) of the MOS devices 202, 204 in order toprevent significant transconductance of the MOS device 202, 204 and theresulting noise. Accordingly, the MOS devices 202, 204 remain in amoderate or weak inversion state when the V_(gs) remains sub-threshold(V_(th)).

As shown, the oscillation frequency may rise as high as a point 310,representing a peak V_(gs) value slightly below V_(th), in a regionlabeled “sub V_(th).” At the point 310, the MOS device 202, 204 mayproduct a small amount of transconductance, and thus a low to moderateamount of thermal noise. However any amount of noise generated may stillbe minimal and centered at the oscillation peaks.

In some embodiments the oscillator 200 is highly sensitive to noise(e.g., phase noise, jitter) as the output voltage (e.g., V_(tank))crosses a value of zero at a node. A line 304 (dashed line) approximatesthe V_(gs) at the nodes, or zero crossing of the oscillator 200 output.The voltage V_(gs) at the line 304 is a negative value, resulting inhigh impedance at the source and drain of the MOS device 202, 204 andcorrespondingly low noise contribution. However, since the thermal noiseat the point 310 occurs at the peak voltage, the phase noisecontribution to the oscillator output is negligible.

The NMOS or PMOS components implemented as the MOS device 202 contributevery little to noise as they spend approximately one quarter of theoscillation cycle in the sub-threshold region (e.g., moderate inversion)and approximately three quarters of the oscillation cycle in thedeep-sub-threshold region (e.g., weak-inversion) where the noise outputof the MOS devices 202 is much lower.

Over time, as the V_(gs) level decreases from the point 310 to a point320, the V_(gs) decreases into a negatively biased region below zero(e.g., a negative V_(gs) voltage), referred to as the “deep sub V_(th),”as noted. In the negatively biased region, the transconductance is zeroor negative, producing a very high impedance and very low thermal noise.

FIG. 4A is a circuit diagram of another embodiment of the oscillator forFIG. 2. As shown, a VCO 400 may be similar to a portion of theoscillator 200 (FIG. 2). The VCO 400 may have the MOS devices 202 as inFIG. 2, however the V_(g) _(_) _(bias) 210 and V_(bias) 120 (FIG. 1) maybe equal (V_(bias)=V_(g) _(_) _(bias) bias) and thus are combined into asingle voltage source V_(bias) at an input V_(bias) 410. In certainembodiments, it is possible to couple the gate and drain of each of theMOS devices 202 to the same voltage source V_(bias) 410. This mayprovide additional options and provide greater simplicity in physicaloscillator circuit design. In some embodiments, the VCO 400 can providesimilar output as the oscillator 200.

In some embodiments of the VCO 400, the bias node of the varactors 112has almost as much voltage swing as the LC tank circuit 110. Thisvoltage swing allows for an active device (e.g., the MOS devices 202) totraverse between moderate-inversion (sub-threshold) to weak inversion(deep sub-threshold) or even to an off state, increasing impedance.

FIG. 4B is comparison of four plot diagrams of voltage, current, andimpedance values of the oscillator of FIG. 4A over time. A plot 450depicts V_(gs) in volts (V) of a MOS device implemented in the VCO 400,(e.g., the MOS devices 202) as a function of time (t) in picoseconds(ps). The plot 450 depicts voltage on the vertical (y) axis versus time(t) on the horizontal (x) axis.

A plot 460 depicts current from drain to source (I_(ds)) in microamps(μA), as a function of time. The plot 460 depicts current on the y-axisversus time (t) on the x-axis.

A plot 470 depicts a variation of voltage across the varactors 112,referred to herein as “V_(tank),” as a function of time (t). TheV_(tank) may be similar to the V 220 tank (FIG. 2). The plot 470 depictsV_(tank) on the y-axis versus time (t) on the x-axis.

A plot 480 depicts the equivalent impedance (Z_(eq)) of one of the MOSdevices used to bias the varactors (e.g., the MOS devices 202, 204) as afunction of time (t). The plot 480 shows the impedance (Z_(eq)) of theMOS devices 202, 204 in ohms (Ω) on the y-axis versus time on thex-axis.

FIG. 4B shows each of the four plot diagrams 450, 460, 470, 480 with thesame time scale where V_(g) _(_) _(bias) is equal to V_(bias). Ingeneral, FIG. 4B depicts measurements taken from the VCO 400 (FIG. 4A)as a function of time.

In an embodiment, the V_(bias) 410 may be set such that the V_(gs) ofthe MOS devices 202 is maintained below the V_(th). In the embodimentdescribed by the plot 450, the maximum V_(gs) attained is approximately0.24V. For example, the V_(th) of the MOS device 202 shown may be 0.25V;therefore the associated source-drain current path is never fully open.This is depicted by the parallel plot 460 of I_(ds) as a function oftime. The current (I_(ds)) is the current flowing from the drain tosource (e.g., trans conductance) of the MOS devices 202 over time,according to the V_(gs) of the MOS device 202 being measured. The I_(ds)varies from a negative value (e.g., drain to source) of approximately−44 μA when the V_(gs) is approximately −0.3V, to a high value ofapproximately +34 μA when the V_(gs) is −0.25V. The I_(ds) currentremains very small, in terms of μA because the V_(gs) remains belowV_(th), minimizing transconductance. In general, the I_(ds) has a delayor lags behind the V_(gs) in time. The lag may vary with outputfrequency and MOS device 202, 204 composition; however, the time lag ofthe current I_(ds) may also produce various spikes in equivalentimpedance (Z_(eq)). One such spike is shown at a point 488 of the plot480, corresponding to the I_(ds) of the MOS device 202 as it inverts(e.g., negative current to positive current) at a point 462 of the plot460.

The plot 470 is shown indicating the zero crossing of V_(tank) 220 at apoint 482. The point 482 coincides with a high Z_(eq) value ofapproximately 10⁴ ohms, or 10 kΩ at a point 484. This preventstransconductance of the MOS device 202 while minimizing phase noise. Asnoted previously, the zero crossing is the point at which an oscillator(e.g., the VCO 400) is most sensitive to phase noise. An increase inphase noise near the zero crossing of the V_(tank) may adversely affectthe oscillator frequency, and therefore the Q factor and spectralpurity. Accordingly, phase noise should be kept to a minimum near thezero crossing. This may be accomplished by maintaining a high Z_(eq) atthe zero crossing as shown. Additionally, maintaining a V_(gs) in asubthreshold region also results in a relatively high average equivalentimpedance (Z_(eq)) as approximated by a dashed line 486.

FIG. 5A is a circuit diagram of another embodiment of the oscillator forFIG. 2. As shown, a DCO 500 may resemble a portion of the oscillator 200(FIG. 2). The DCO 500 may have the MOS devices 204 as in FIG. 2, howeverthe V_(g) _(_) _(bias2) 222 and the band control 140 (FIG. 1) may beequal (V_(g) _(_) _(bias2) band_control) and thus are combined into asingle voltage source labeled band control 510. In certain embodiments,the gate and drain of each of the MOS devices 204 may be coupled to thesame input to receive an equivalent voltage source from a band control510. This may provide additional options and provide greater simplicityin physical oscillator circuit design. In some embodiments, the DCO 500is similar to the oscillator 200.

In an embodiment, the DCO 500 may be configured to provide an outputV_(tank3) 520 a and V_(tank) 520 b, collectively referred to as“V_(tank) 520.” The V_(tank) 520 may be a digital clock signal orsimilar DCO 500 output.

FIG. 5B is a comparison of four plot diagrams of voltage, current, andimpedance values of the oscillator circuit of FIG. 5A over time. A plot550 depicts V_(gs) in volts (V) of a MOS device implemented in the DCO500, (e.g., the MOS devices 204) as a function of time (t) inpicoseconds (ps). The plot 550 depicts voltage on the vertical (y) axisversus time (t) on the horizontal (x) axis.

A plot 550 depicts V_(gs) in volts (V) of a MOS device implemented inthe DCO 500, (e.g., the MOS devices 204) as a function of time (t) inpicoseconds (ps). The plot 550 depicts voltage on the vertical (y) axisversus time (t) on the horizontal (x) axis.

A plot 560 depicts current from drain to source (I_(ds)) in microamps(μA), as a function of time. The plot 460 depicts current on the y-axisversus time (t) on the x-axis.

A plot 570 depicts the equivalent impedance (Z_(eq)) of one of the MOSdevices used to bias the DCO 500 (e.g., the MOS devices 204) as afunction of time (t). The plot 570 shows the impedance (Z_(eq)) of theMOS devices 204 in ohms (Ω) on the y-axis versus time on the x-axis.

A plot 580 depicts a variation of voltage output pf the DCO 500,referred to herein as “V_(tank),” as a function of time (t). TheV_(tank) may be similar to the V 220 tank (FIG. 2) but a DCO output asopposed to a VCO output. The plot 580 depicts V_(tank) on the y-axisversus time (t) on the x-axis.

FIG. 5B shows each of the four plot diagrams 550, 560, 570, 580 with thesame time scale where V_(g) _(_) _(bias2) 222 (FIG. 2) is equal to bandcontrol 140 (FIG. 2). In general, FIG. 5B depicts oscillator voltagetaken from the DCO 500 (FIG. 5A) output as a function of time.

In an embodiment, the band control 510 may be set such that the V_(gs)of the MOS devices 204 is maintained below the V_(th). In the embodimentdescribed by the plot 550, the maximum V_(gs) attained is approximately0.24V, 0r 240 millivolts (mV), as shown. For example, the V_(th) of theMOS device 202 may be 250 mV, therefore the source-drain (I_(ds))current path is never fully open allowing only a minimum current. Thisis depicted by the parallel plot 460 of I_(ds) as a function of time.The current (I_(ds)) is the current flowing from the drain to source(e.g., transconductance) of the MOS device 204 over time, according tothe V_(gs) of the same MOS device 204. The I_(ds) varies from a positivevalue (e.g., drain to source) of approximately +3.5 μA, to a value ofapproximately −3 μA. Similar to FIG. 4B, the I_(ds) current remains verysmall, in terms of μA because the V_(gs) remains below V_(th),minimizing transconductance and maintaining high impedance. In general,the I_(ds) is delayed behind the V_(gs) in time. The lag may vary withDCO 500 output frequency, however, the time lag of the current I_(ds)may also produce various spikes in equivalent impedance Z_(eq), (shownin the plot 570) as the I_(ds) of the MOS device 204 inverts at a point562. The point 562 thus corresponds with an spike in Z_(eq), at a point588 as the I_(ds) inverts at the point 562. This may be a similarphenomenon to that shown in FIG. 4B and the plot 480.

The plot 580 is shown indicating the zero crossing of V_(tank) 520 at apoint 582. The point 582 coincides with a Z_(eq) value of approximately10⁵ ohms, or 100 kΩ at a point 584 on the plot 570. This high impedanceprevents transconductance of the MOS device 204 while minimizing phasenoise or jitter. As noted previously, the zero crossing is the point atwhich a digitally controlled oscillator (e.g., the 500) is mostsensitive to jitter. An increase in jitter near the zero crossing of theV_(tank) 520 may adversely affect the frequency, accuracy, and precisionof the DCO 500. Accordingly, jitter should be kept to a minimum near thezero crossing. This may be accomplished by maintaining a high Z_(eq) atthe zero crossing as shown. Additionally, maintaining a V_(gs) in asubthreshold region also results in relatively high average equivalentimpedance (Z_(eq)) as approximated by a dashed line 586.

When one of the coarse capacitors C₃ 114, C₄ 116, is switched into theDCO 500, the voltage swing at the MOS devices 204 is very low on thisnode. The long/narrow biasing NMOS provides a high impedance, indicatedby the plot 570. When the other coarse capacitor (C₃ 114, C₄ 116) isswitched out of the DCO 500, the voltage swing drives high, allowing forthe MOS devices 204 to traverse between moderate-inversion(sub-threshold) to weak inversion (deep sub-threshold) as shown in theplot 300 (FIG. 3). The deep subthreshold region may also produce an“off” state, eliminating current flow (I_(ds)) and again providing thedesired high impedance (Z_(eq)).

The above description of the disclosed embodiment is provided to enableany person skilled in the art to make or use the disclosure. Variousmodifications to these embodiments will be readily apparent to thoseskilled in the art, and the generic principles described herein can beapplied to other embodiment without departing from the spirit or scopeof the disclosure. Thus, it is to be understood that the description anddrawings presented herein represent a presently preferred implementationof the disclosure and are therefore representative of the subject matterwhich is broadly contemplated by the present disclosure. It is furtherunderstood that the scope of the present disclosure fully encompassesother embodiment that may become obvious to those skilled in the art andthat the scope of the present disclosure is accordingly limited bynothing other than the appended claims.

What is claimed is:
 1. A frequency oscillator comprising: a tank circuithaving an inductor, a first coupling capacitor, and a second couplingcapacitor; a varactor circuit electrically coupled to the first couplingcapacitor and the second coupling capacitor; a first MOS device having afirst gate, a first drain, and a first source, the first source beingelectrically coupled to the varactor circuit, and the first gate beingelectrically coupled to the first drain; a second MOS device having asecond gate, a second drain, and a second source, the second sourcebeing electrically coupled to the varactor circuit opposite the firstsource and the second gate being electrically coupled to the seconddrain; a first input electrically coupled to the first drain and thesecond drain operable to receive a first bias voltage; and a secondinput electrically coupled to the first gate and the second gate toreceive a first gate bias voltage.
 2. The frequency oscillator of claim1 further comprising: a first coarse tuning capacitor electricallycoupled to the first coupling capacitor and the inductor; a secondcoarse tuning capacitor electrically coupled to the second couplingcapacitor and the inductor; a third MOS device having a third gate, athird drain, and a third source, the third drain being electricallycoupled to the first coarse tuning capacitor; a fourth MOS device havingfourth gate, a fourth drain, and a fourth source, the fourth drain beingelectrically coupled to the second coarse tuning capacitor; a bandcontrol input electrically coupled to the third source and the fourthsource; a third input electrically coupled to the third gate and thefourth gate to receive a second gate bias voltage.
 3. The frequencyoscillator of claim 2, wherein the band control input is electricallycoupled to the third input.
 4. The frequency oscillator of claim 2,wherein the band control input and the third input are configured toreceive at least one bias voltage, the at least one bias voltageselected to negatively bias the third MOS device and the fourth MOSdevice.
 5. The frequency oscillator of claim 2 further comprising aswitch having a switch gate, a switch drain, and a switch source, theswitch gate being electrically coupled to the band control input, theswitch drain being electrically coupled to the first coarse tuningcapacitor and the third MOS device, and the switch source beingelectrically coupled to the second coarse tuning capacitor and thefourth MOS device.
 6. The frequency oscillator of claim 1, wherein thefirst input is electrically coupled to the second input, and wherein thefirst gate bias voltage is equal to the first bias voltage.
 7. Thefrequency oscillator of claim 1, wherein the first input and the secondinput are configured to receive at least one bias voltage, the at leastone bias voltage selected to negatively bias the first MOS device andthe second MOS device.
 8. The frequency oscillator of claim 1, whereinthe first MOS device and the second MOS device are configured to biasthe varactor circuit and isolate a phase noise contribution of the firstMOS device and the second MOS device to an oscillation peak of an outputof the frequency oscillator.
 9. A frequency oscillator comprising: avariable capacitance circuit; a tank circuit having at least oneinductor and at least one capacitor, the tank circuit being electricallycoupled in parallel to the variable capacitance circuit; a first MOSdevice having a first gate, a first source, and a first drain, the firstsource electrically coupled to the tank circuit and the variablecapacitance circuit, and the first gate being electrically coupled tothe first drain; a second MOS device having a second gate, a secondsource, and a second drain, the second source electrically coupled tothe tank circuit and the variable capacitance circuit, and the secondgate being electrically coupled to the second drain; a first inputelectrically coupled to the first drain and the second drain, andconfigured to receive a first bias voltage; and a second inputelectrically coupled to the first gate and the second gate, the secondinput configured to receive a first gate bias voltage, the first gatebias voltage operable to bias the first MOS device such that a firstgate-to-source voltage of the first MOS device remains below a firstthreshold voltage, and to bias the second MOS device such that a secondgate-to-source voltage of the second MOS device remains below a secondthreshold voltage, when the frequency oscillator is in operation. 10.The frequency oscillator of claim 9, further comprising: a first coarsetuning capacitor and a second coarse tuning capacitor electricallycoupled to the tank circuit; a third MOS device having a third gate, athird source, and a third drain, the third source electrically coupledto the first coarse tuning capacitor; a fourth MOS device having afourth gate, a fourth source, and a fourth drain, the fourth sourceelectrically coupled to the second coarse tuning capacitor; a bandcontrol input electrically coupled to the third drain and the fourthdrain; and a third input electrically coupled to the third gate and thefourth gate to receive a second gate bias voltage, the second gate biasvoltage being configured to bias the third MOS device such that a thirdgate-to-source voltage remains below a third threshold voltage and tobias the fourth MOS device such that a fourth gate-to-source voltageremains below a fourth threshold voltage.
 11. The frequency oscillatorof claim 10, wherein the band control input is electrically coupled tothe third input.
 12. The frequency oscillator of claim 9, wherein thefirst input is electrically coupled to the second input, and wherein thefirst gate bias voltage is equal to the first bias voltage.
 13. Thefrequency oscillator of claim 9, wherein the first input and the secondinput are configured to receive at least one bias voltage, the at leastone bias voltage selected to negatively bias the first MOS device andthe second MOS device.
 14. A method for biasing an oscillator circuit,comprising: generating an oscillating output using a tank circuitelectrically coupled to a varactor circuit; biasing the varactor circuitusing a first MOS device having a first threshold voltage and a secondMOS device having a second threshold voltage, the varactor circuit beingelectrically coupled to a first source of the first MOS device and to asecond source of the second MOS device; biasing the first MOS device andthe second MOS device with a first gate bias voltage at a first gate ofthe first MOS device and at a second gate of the second MOS device;electrically coupling the first gate of the first MOS device to a firstdrain of the first MOS device; electrically coupling the second gate ofthe second MOS device to a second drain of the second MOS device; andcontrolling a first transconductance of the first MOS device and asecond transconductance of the second MOS device with a first biasvoltage and the first gate bias voltage.
 15. The method of claim 14,further comprising: generating a digital clock signal using the tankcircuit and a band control input, the tank circuit being electricallycoupled to a first coarse tuning capacitor and a second coarse tuningcapacitor; biasing a third MOS device and a fourth MOS device with asecond gate bias voltage at a third gate of the third MOS device and ata fourth gate of the fourth MOS device, the band control input beingelectrically coupled to a third source of the third MOS device and afourth source of the fourth MOS device; and controlling a thirdtransconductance of the third MOS device and a fourth transconductanceof the fourth MOS device using the band control input and the secondgate bias voltage.
 16. The method of claim 15 further comprisingelectrically coupling the band control input to the second gate biasvoltage.
 17. The method of claim 14, further comprising electricallycoupling the first gate bias voltage to the first bias voltage.
 18. Themethod of claim 14, wherein a difference between the first bias voltageand the first gate bias voltage is not greater than the first and secondthreshold voltages.
 19. An apparatus for producing an oscillatingfrequency comprising: a resonating means for storing energy at aresonant frequency, the resonating means having and at least oneinductor and at least one capacitor; a variable capacitance means havinga first end and a second end, the first end and the second end beingelectrically coupled to the resonating means; a first transistor meanshaving a first gate, a first drain, and a first source, the first sourcebeing electrically coupled to the first end, and the first gate beingelectrically coupled to the first drain; a second transistor meanshaving a second gate, a second drain, and a second source, the secondsource being electrically coupled to the second end, and the second gatebeing electrically coupled to the second drain; a first biasing meanselectrically coupled to the first drain and the second drain; and asecond biasing means electrically coupled to the first gate and thesecond gate.
 20. The apparatus of claim 19, wherein the variablecapacitance means comprises a varactor circuit having the first end andthe second end, the first end being electrically coupled to a firstcoupling capacitor and the second end being electrically coupled to asecond coupling capacitor.
 21. The apparatus of claim 19, furthercomprising: a third transistor means having a third gate, a thirdsource, and a third drain, the third source electrically coupled to theresonating means; a fourth transistor means having a fourth gate, afourth source, and a fourth drain, the fourth source being electricallycoupled to the resonating means; a band control means electricallycoupled to the third drain and the fourth drain; and an input meanselectrically coupled to the third gate and the fourth gate to receive athird biasing means.
 22. The apparatus of claim 21, wherein the thirdbiasing means is configured to bias the third transistor means such thata third gate-to-source voltage remains below a third threshold voltageand to bias the fourth transistor means such that a fourthgate-to-source voltage remains below a fourth threshold voltage.
 23. Theapparatus of claim 21 further comprising a switching means having aswitch source, a switch drain, and a switch gate, the switch gate beingelectrically coupled to the band control means, the switch drain beingelectrically coupled to the third transistor means and the resonatingmeans, and the switch source being electrically coupled to the fourthtransistor means and the resonating means.